Anritsu HFE0503_Leong 电路图.pdf

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1、42High Frequency Electronics High Frequency Design BROADBAND DESIGN Broadband VHF/UHF Amplifier Design Using Coaxial Transformers By C. G. Gentzler and S.K. Leong Polyfet RF Devices T he desire of the armed forces to maintain instant communications with all forces requires the design of miniature br

2、oadband power amplifiers with greater than decade bandwidth (30 to 512 MHz). This bandwidth is required for all-band transceivers that cover tactical ground and air frequencies in addition to civil telecommunication frequencies and the frequencies of our allies. All-band radios are commonplace in vi

3、rtually every deployment of new platforms, as well as in the retrofitting of existing communications systems. This paper will discuss the design of minia- ture coaxial structures and examine the implementation of improved design tech- niques to enable the designer to obtain insight into matching the

4、 load line of power MOSFET transistors over decade bandwidths. This article presents the development of large signal parameters for a typical power MOSFET device and the development of a suitable load line using coaxial transmission line transformers in conjunction with embed- ded lumped structures,

5、 enabling an efficient load line match across a decade of bandwidth. Simulation Methodology. Linear simulation assumes that a circuit with active devices is operated at such a low power that the simulated measurements are no longer power dependent. This simulation can be achieved by two methods. Fir

6、st the cir- cuit uses a nonlinear model and nonlinear simulator. The quiescent current is set at a nominal condition and the power level used in the simulation is set to a low level so as not to make the output data power dependent. Another linear simulation method is to use tabular data to describe

7、 an active device and simulate with a linear simulator. Usually the data file is in S-parameter format although other formats have been used in the past at lower frequencies (e.g. impedance magnitude and angle). If the nonlinear model and the data file agree, both simulations will yield the same mea

8、surement data. In the case of using a non-linear model with a nonlinear simula- tor, the simulation results are generally very close to actual amplifier performance. Nonlinear simulators provide gain com- pression, power output, efficiency and har- monic power data. With somewhat less accu- racy, in

9、termodulation distortion can be simu- lated, but not with the same accuracy as the single tone measurements. To obtain accurate results, the device model would have to track an actual device transfer curve closer than 5 percent. Five percent accuracy is generally acceptable for gain compression and

10、efficiency measurements, but not for the slight nonlin- earity that causes low to intermediate levels of intermodulation distortion. Modeling tech- nology is slowing improving and it is expected that intermodulation performance may be accurately modeled in the future. Nonlinear simulators generally

11、are more costly, but are really the only choice if large signal perfor- mance simulation is desired, as in the case of this article. Amplifier Design First one must determine the optimum load line impedance required by the device. This article describes the methods used to design broadband coaxial t

12、rans- former matching networks for an LDMOS power ampli- fier that delivers consistent performance over more than a decade bandwidth From May 2003 High Frequency Electronics Copyright 2003 Summit Technical Media, LLC RadioFans.CN 收音机爱 好者资料库 44High Frequency Electronics High Frequency Design BROADBAN

13、D DESIGN Computer load pull or optimization is required since any actual load pull techniques are only generally avail- able for much higher frequencies.The physical structures for generating load impedances at frequencies below 500 MHz are too large to be practical to implement. Additionally, since

14、 the band width is multi-octave, broad- band matching structures must be used to determine the load line rather than multiple narrow band measurements.A computer with suit- able software and good device models is the most practical approach. In this article we will use popular soft- ware packages su

15、ch as Applied Wave Researchs (AWR) Microwave Office and Agilent Technologies ADS, used together with Polyfet RF Spice Models to demonstrate broadband matching techniques. Impedance Behavior of Transistors At low frequencies, the devices output impedance is relatively high compared with the calculate

16、d load line required to produce the desired power. As the operating frequency is increased, the output capacitance (Coss), reverse capacitance (Crss) and an increased saturation voltage low- ers the optimum load line to achieve satisfactory performance. Over a decade of bandwidth, the optimum impeda

17、nce can drop by a factor of two.That is to say that if the low frequency load line is 6 ohms, the upper operating frequency could require an impedance of 3 ohms with some amount of inductive or capaci- tive reactance. Figure 1 shows real value of Zoutdropping from 11 ohms at low frequencies to 2 ohm

18、s at high frequency for the transistor LR401. There has been considerable experimental and developmental work published on the attributes of coaxial transformers to achieve extremely wide bandwidths. This paper will explore how to combine the coaxial transformer with lumped components to achieve opt

19、imal power matching in a MOSFET power ampli- fier over more than a decade of band- width. Computer simulated load pulling will be utilized to extract the first order magnitude of load line match- ing. This impedance information is only the starting point, since it will be extracted by a narrow band

20、tech- nique. Broadband extraction is an area that will be explored in the future as the results will take into account more realistic harmonic load- ing and allow more accurate broad- band design implementation. In the case of Polyfet transistors, Zin/Zout data can be found for each transistor in it

21、s respective data sheet. Once the approximate load line has been determined, let us review the coaxial transformer matching techniques and explore the use of physical length, cable impedance, and lumped components in addition to ferrite loading to achieve optimum performance. Of all the coaxial tran

22、sformer designs, one of the most practical for wideband impedance matching is the 4:1 design with a balun transformer to achieve optimum balance. The standard accepted equation for trans- formation is that the Z0of the cable should be the square root of the prod- uct of the input and output impedanc

23、es. For example, if the input impedance is 12.5 ohms and the out- put impedance is 50 ohms, then the square root of 12.5 50 = 25. A 25- ohm impedance cable would give the optimum results across a wide oper- ating bandwidth. Figure 3 shows a uniform impedance transformation ratio of four across the f

24、requency band. It should be noted for the purpose of load line design, impedance is mea- sured drain to drain. This allows sin- gle ended impedance measurements. Simply divide the impedance data by two to obtain individual device load impedance. At 30 MHz the ratio falls off due to reactive shunt lo

25、sses, which could be compensated with Figure 1 Zinand Zout vs. frequency.Figure 2 Conventional 4:1 transformer with balun. RadioFans.CN 收音机爱 好者资料库 46High Frequency Electronics High Frequency Design BROADBAND DESIGN ferrite loading. The object is to design a load line that lowers the real resistance

26、as the frequency increases.This requires some rethinking as to how one might exploit the benefits of transmission line matching in conjunction with techniques mentioned above to achieve a satisfactory load line over a decade of bandwidth. A Novel Approach The following is a presentation of how to em

27、bed a lumped matching network into a transmission match- ing network to achieve a suitable broad band power match. A conven- tional design allows the coaxial transformer to transform the impedance to obtain a match the low end of the band, then add additional lowpass matching sections to lower the i

28、mpedance at the upper band edge. Although this technique per- forms satisfactorily,a microstrip implementation would occupy consid- erable space. A novel approach to this problem, shown in Figure 4, is to use the effec- tive inductance of the coaxial trans- mission lines as the inductive compo- nent

29、 in a pi matching network. Only small chip capacitors will be needed to complete the transformation at the upper band edge. By a selection of the transmission line impedance and electrical length, a load line may be created that will essentially provide the basic transformation ratios at the lower b

30、and edge. As the operating frequency is increased the combina- tion of the transmission line effective inductance and the shunt capaci- tance will lower the load line to effec- tively match the device at the upper band edge.Figure 5 shows the impedance dropping with increasing frequency. This can be

31、 accomplished with the same physical constraints as just a broadband transformer alone. This technique enables one to con- struct decade bandwidth power amplifiers with physical dimensions no larger than the transformers and the device. The savings in size can be critical in some applications. Desig

32、ning the Load Line To design our example 80 watt broadband amplifier that covers 30 512 MHz band, one would first calcu- late the load line for the lower band edge. Using a simple approximation of Steve Cripps law 1, lets calculate the low frequency load line. (285)2/(280) = 3.31 ohms or 6.62 ohms f

33、or two push-pull devices. A 6.25-ohm load line that is achieved with a 4:1 coaxial transformer and a 1:1 balun easily accomplishes this task. Next, using simulator generated R VV P load ddsat out = () 2 2 Frequency Coax transformer (GHz) Zin1 Zin1 (Real)(Imag.) 0.0311.104.92 0.0812.941.91 0.13 13.07

34、 0.94 0.18 12.99 0.45 0.23 12.87 0.17 0.28 12.73 0.02 0.33 12.60 0.05 0.38 12.49 0.06 0.43 12.41 0.04 0.48 12.37 0.00 0.53 12.36 0.05 0.58 12.37 0.08 0.6 12.38 0.09 Figure 3 Uniform 4:1 transformation across the frequency band.Figure 4 Variable 4:1 impedance transformer and matching network. Frequen

35、cy Coax transformer (GHz) Zin1 Zin1 (Real)(Imag.) 0.0311.914.13 0.0812.171.02 0.1310.062.17 0.188.351.77 0.237.290.79 0.28 6.75 0.34 0.33 6.64 1.42 0.38 6.83 2.33 0.43 7.21 2.95 0.48 7.62 3.22 0.53 7.86 3.20 0.58 7.80 3.05 0.6 7.69 2.99 Figure 5 Impedance decreases with frequency. RadioFans.CN 收音机爱

36、好者资料库 48High Frequency Electronics High Frequency Design BROADBAND DESIGN large signal impedance data, review the optimum match at the upper band edge. The next step is to use a linear simulator to embed the match- ing structure into the transformer structure. In order to successfully embed the uppe

37、r edge matching net- work into the transformer the electri- cal length of the transformer should be shorter that 1/8 wavelength at the highest operating frequency.This will keep the transmission line acting as an inductance. Both the length and impedance may be varied to optimize the performance ove

38、r the band. For example, to design an embed- ded matching network for the Polyfet LR401 push-pull MOSFET device, one would start with the power level desired and determine the low fre- quency load line. Since the low fre- quency load line is output power related and not necessarily a function of the

39、 output impedance of the device, we will use a 4:1 coaxial transformer followed by a 1:1 balun transform to establish a solid 6.25 ohm load line from the lower band edge up to several octaves higher or around 120 MHz. Above 120 MHz, the large signal impedance will determine the impedance transfor- m

40、ation required to maintain ade- quate performance. The technique in broadband matching is normally to match the highest frequency and use the fact the power impedance con- tours where satisfactory operation can be obtained become larger as the operating frequency is reduced. Large Signal Simulation

41、Once the load line has been designed, it is time for large signal simulation. The input matching sec- tion is designed in a similar manner as the output section with the excep- tion that since the return loss can be measured during simulation, it is much easier to either manually tune or automatical

42、ly optimize the input circuit. Assuming the tentative circuit design has been completed, the next step is nonlinear simulation. It is strongly recommended to start the simulation at a low input power level and check for small signal gain, gain flatness, and input return loss. The input return loss m

43、ay be tuned under small signal conditions since it will not change significantly as the power level is increased. Do not attempt to tune on the output matching section under small signal operation, since the load line tuning is extremely power sensitive. Once satisfactory Figure 6 Input schematic fo

44、r nonlinear simulation. Figure 7 Input circuit layout diagram.Figure 8 Actual amplifier; note the small size. 50High Frequency Electronics High Frequency Design BROADBAND DESIGN gain and return loss has been obtained under small signal condi- tion, slowly raise the input power until the amplifier st

45、arts to com- press. Typically, the compression will first occur at mid to higher frequen- cies.The goal of high power optimiza- tion is to obtain a flat compression point across the highest octave of amplifier operation. Manual tuning or an optimization feature may accomplish this goal. Manual tunin

46、g is usually the best avenue of approach since most optimization routines are somewhat linear simula- tion based, and variables (component values) have to be constrained great- ly in order to get meaningful results. With todays Pentium computers and improved EDA software, nonlinear simulation speeds

47、 approach that of linear simulation just a few years earlier. Real time nonlinear tuning is a capability of present simulators. As the optimum load line is approached, slight optimization of the input cir- cuit will be required to obtain an opti- mum input return loss. Since the out- put load line t

48、uning has minimal effect on the input tuning only a slight adjustment should be required. The circuit used in simulation shown in Figure 6 consists of similar input and output matching networks, as described earlier in this article. The series R-C on the output of the input balun acts as series gate

49、 resis- tance to lower the gain of the transis- tor. A series RLC network between gate and source is added to stabilize the transistor from low frequency oscillations and series RLC drain to gate feedback is added to further enhance stability and achieve a flat gain over the band. The schematic shows additional inductances to rep- resent the printed circuit board pads for component mounting. The drains of the transistors are fed to DC sup- plies through chokes that are repre- sented by air coils. At the DC supply feed, there is a choke with a parallel resistor to further increa