Anritsu HFE0903_RaabPart3 电路图.pdf

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1、34High Frequency Electronics High Frequency Design RF POWER AMPLIFIERS RF and Microwave Power Amplifier and Transmitter Technologies Part 3 By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington, Zoya B. Popovich, Nick Pothecary, John F. Sevic and Nathan O. Sokal T he building blocks u

2、sed in transmit- ters are not only power amplifiers, but a variety of other circuit elements including oscil- lators, mixers, low-level amplifiers, filters, match- ing networks, combiners, and circulators.The arrangement of building blocks is known as the architecture of a transmitter. The classic t

3、ransmitter architecture is based upon linear PAs and power combiners. More recently, transmitters are being based upon a variety of different architectures including stage bypassing, Kahn, envelope tracking, outphas- ing, and Doherty. Many of these are actually fairly old techniques that have been r

4、ecently made practical by the capabilities of DSP. 7a. LINEAR ARCHITECTURE The conventional architecture for a linear microwave transmitter consists of a baseband or IF modulator, an up-converter, and a power- amplifier chain (Figure 20). The amplifier chain consists of cascaded gain stages with pow

5、er gains in the range of 6 to 20 dB. If the transmitter must produce an amplitude-mod- ulated or multi-carrier signal, each stage must have adequate linearity.This generally requires class-A amplifiers with substantial power back-off for all of the driver stages. The final amplifier (output stage) i

6、s always the most costly in terms of device size and current consumption, hence it is desirable to operate the output stage in class B. In applications requiring very high linearity, it is necessary to use class A in spite of the lower efficiency. The outputs of a driver stage must be matched to the

7、 input of the following stage much as the final amplifier is matched to the load. The matching tolerance for maintaining power level can be significantly lower than that for gain 60, hence the 1-dB load-pull contours are more tightly packed for power than for gain. To obtain even modest bandwidths (

8、e.g., above 5 percent), the use of quadrature bal- anced stages is advisable (Figure 21). The main benefit of the quadrature balanced con- figuration is that reflections from the transis- tors are cancelled by the action of the input and output couplers. An individual device can therefore be deliber

9、ately mismatched (e.g., to achieve a power match on the output), yet the quadrature-combined system appears to be well-matched. This configuration also acts as an effective power combiner, so that a given power rating can be achieved using a pair of devices having half of the required power per- for

10、mance. For moderate-bandwidth designs, the lower-power stages are typically designed using a simple single-ended cascade, which in some cases is available as an RFIC. Designs with bandwidths approaching an octave or Transmitter architectures is the subject of this install- ment of our continuing ser

11、ies on power amplifiers, with an emphasis on designs that can meet todays linearity and high efficiency requirements Figure 20 A conventional transmitter. RF/ Baseband Exciter Mixer LO RX 3-stage PA From September 2003 High Frequency Electronics Copyright 2003 Summit Technical Media, LLC RadioFans.C

12、N 收音机爱 好者资料库 36High Frequency Electronics High Frequency Design RF POWER AMPLIFIERS more require the use of quadrature- balanced stages throughout the entire chain. Simple linear-amplifier chains of this kind have high linearity but only modest efficiency.Single-carrier applications usually operate

13、the final amplifier to about the 1-dB compres- sion point on amplitude modulation peaks. A thus-designed chain in which only the output stage exhibits compression can still deliver an ACPR in the range of about 25 dBc with 50-percent efficiency at PEP. Two practical problems are fre- quently encount

14、ered in the design of linear PA chains: stability and low gain. Linear, class-A chains are actu- ally more susceptible to oscillation due to their high gain, and single- path chains are especially prone to unstable behavior. Instability can be subdivided into the two distinct cate- gories: Low-frequ

15、ency oscillation and in-band instability. In-band instabili- ty is avoided by designing the indi- vidual gain stages to meet the crite- ria for unconditional stability; i.e., the Rollet k factor 61 must be greater than unity for both in-band and out-of-band frequencies. Meeting this criterion usuall

16、y requires sacri- ficing some gain through the use of absorptive elements. Alternatively, the use of quadrature balanced stages provides much greater isola- tion between individual stages, and the broadband response of the quadrature couplers can eliminate the need to design the transistor stage its

17、elf with k1. This is another reason for using quadrature coupled stages in the output of the chain. Large RF power devices typically have very high transconductance, and this can produce low-frequency insta- bility unless great care is taken to terminate both the input and output at low frequencies

18、with impedances for unconditional stability. Because of large separation from the RF band, this is usually a simple matter requir- ing a few resistors and capacitors. At X band and higher, the power gain of devices in the 10 W and above category can drop well below 10 dB. To maintain linearity, it m

19、ay be nec- essary to use a similarly size device as a driver. Such an architecture clearly has a major negative impact upon the cost and efficiency of the whole chain. In the more extreme cases, it may be advantageous to con- sider a multi-way power combiner, where 4, 8, or an even greater num- ber

20、of smaller devices are combined. Such an approach also has other advantages, such as soft failure, bet- ter thermal management, and phase linearity. However, it typically con- sumes more board space. 7b. POWER COMBINERS The need frequently arises to combine the outputs of several indi- vidual PAs to

21、 achieve the desired transmitter output. Whether to use a number of smaller PAs vs. a single larger PA is one of the most basic decisions in selection of an architec- ture 60. Even when larger devices are available, smaller devices often offer higher gain, a lower matching Q factor (wider bandwidth)

22、,better phase linearity, and lower cost. Heat dissipation is more readily accom- plished with a number of small devices, and a soft-failure mode becomes possible. On the other hand, the increase in parts count, assembly time, and physical size are significant disadvantages to the use of multiple, sm

23、aller devices. Direct connection of multiple PAs is generally impractical as the PAs interact, allowing changes in output from one to cause the load impedance seen by the other to vary. A constant load impedance, hence isolation of one PA from the other, is provided by a hybrid combiner.A hybrid com

24、biner causes the difference between the two PA outputs to be routed to and dissipated in a balancing or “dump” resistor. In the event that one PA fails, the other continues to operate normally, with the transmitter out- put reduced to one fourth of nominal. The most common power combin- er is the qu

25、adrature-hybrid combiner. A 90 phase shift is introduced at input of one PA and also at the out- put of the other. The benefits of quadrature combining include con- stant input impedance in spite of variations of input impedances of the individual PAs, cancellation of odd harmonics, and cancellation

26、 of back- ward-IMD (IMD resulting from a sig- nal entering the output port). In addition, the effect of load impedance upon the system output is greatly reduced (e.g., to 1.2 dB for a 3:1 SWR). Maintenance of a nearly con- stant output occurs because the load impedance presented to one PA decreases

27、when that presented to the other PA increases. As a result, how- ever, device ratings increase and effi- ciency decreases roughly in propor- tion to the SWR 65.Because quadrature combiners are inherently two-terminal devices, they are used in a corporate combining architecture Figure 21 Amplifier st

28、ages with quadrature combiners. 0 90 RadioFans.CN 收音机爱 好者资料库 38High Frequency Electronics High Frequency Design RF POWER AMPLIFIERS (Figure 21). Unfortunately, the physical construction of such couplers poses some problems in a PC-board envi- ronment. The very tight coupling between the two quar- te

29、r-wave transmission lines requires either very fine gaps or a three-dimensional structure. This problem is circum- vented by the use of a miniature co-axial cable having a pair of precisely twisted wires to from the coupling sec- tion or ready-made, low-cost surface mount 3-dB couplers. The Wilkinso

30、n or in-phase power combiner 62 is often more easily fabricated than a quadrature combiner. In the two-input form (as in each section in Figure 22), the outputs from two quarter-wavelength lines summed into load R0produce an apparent load impedance of 2R0, which is transformed through the lines into

31、 at the load impedances RPAseen by the individual PAs. The differ- ence between the two PA outputs is dissipated in a resis- tor connected across the two inputs. Proper choice of the balancing resistor (2RPA) produces a hybrid combiner with good isolation between the two PAs. The Wilkinson concept c

32、an be extended to include more than two inputs 63. Greater bandwidth can be obtained by increasing the number of transforming sections in each signal path. A single-section combiner can have a useful bandwidth of about 20 percent, whereas a two-section version can have a bandwidth close to an octave

33、. In practice, escalating cir- cuit losses generally preclude the use of more than two sections. All power-combining techniques all suffer from circuit losses as well as mismatch losses. The losses in a simple two-way combiner are typically about 0.5 dB or 10 per- cent. For a four-way corporate stru

34、cture, the intercon- nects typically result in higher losses. Simple open microstrip lines are too lossy for use in combining struc- tures. One technique that offers a good compromise among cost, produceability, and performance, uses sus- pended stripline. The conductors are etched onto double- side

35、d PC board, interconnected by vias, and then sus- pended in a machined cavity. Structures of this kind allow high-power 8-way combiners with octave bandwidths and of 0.5 dB. A wide variety of other approaches to power-combin- ing circuits are possible 62, 64. Microwave power can also be combined dur

36、ing radiation from multiple anten- nas through “quasi-optical” techniques 66. 7c. STAGE SWITCHING AND BYPASSING The power amplifier in a portable transmitter gener- ally operates well below PEP output, as discussed in Section 4 (Part 1). The size of the transistor, quiescent current, and supply volt

37、age are, however, determined by the peak output of the PA. Consequently, a PA with a lower peak output produces low-amplitude signals more efficiently than does a PA with a larger peak output, as illustrated in Figure 23 for class-B PAs with PEP effi- ciencies of 60 percent. Stage-bypassing and gate

38、-switch- ing techniques 67, 68 reduce power consumption and increase efficiency by switching between large and small amplifiers according to signal level. This process is analo- gous to selection of supply voltage in a class-G PA, and the average efficiency can be similarly computed 69. A typical st

39、age-bypassing architecture is shown in Figure 24. For low-power operation, switches SA and SB route the drive signal around the final amplifier. Figure 22 Multi-section Wilkinson combining architecture. Figure 23 Power consumption by PAs of different sizes. Figure 24 Stage-bypassing architecture. Ra

40、dioFans.CN 收音机爱 好者资料库 40High Frequency Electronics High Frequency Design RF POWER AMPLIFIERS Simultaneously, switch SDC turns-off DC power to the final amplifier. The reduction in power consumption can improve the average efficiency signif- icantly (e.g., from 2.1 to 9.5 percent in 70).The control s

41、ignal is based upon the signal envelope and power level (back-off). Avoiding hysteresis effects and distortion due to switching tran- sients are critical issues in imple- mentation. A PA with adaptive gate switching is shown in Figure 25.The gate width (hence current and power capability) of the upp

42、er FET is typically ten to twenty times that of the lower FET. The gate bias for the high-power FET keeps it turned off unless it is needed to support a high-power output. Consequently, the quiescent drain current is reduced to low levels unless actually needed. The advantages of this technique are

43、the absence of loss in the switches required by stage bypassing, and operation of the low- power FET in a more linear region (vs. varying the gate bias of a single large FET). The disadvantage is that the source and load impedances change as the upper FET is switched on and off. 7d. KAHN TECHNIQUE T

44、he Kahn Envelope Elimination and Restoration (EER) technique (Figure 26) combines a highly effi- cient but nonlinear RF power amplifi- er (PA) with a highly efficient enve- lope amplifier to implement a high- efficiency linear RF power amplifier. In its classic form 73, a limiter elim- inates the en

45、velope, allowing the con- stant-amplitude phase modulated carrier to be amplified efficiently by class-C,-D,-E,or -F RF PAs. Amplitude modulation of the final RF PA restores the envelope to the phase- modulated carrier creating an ampli- fied replica of the input signal. EER is based upon the equiva

46、- lence of any narrowband signal to simultaneous amplitude (envelope) and phase modulations. In a modern implementation, both the envelope and the phase-modulated carrier are generated by a DSP. In contrast to linear amplifiers, a Kahn-technique transmitter operates with high effi- ciency over a wid

47、e dynamic range and therefore produces a high aver- age efficiency for a wide range of sig- nals and power (back-off) levels. Average efficiencies three to five times those of linear amplifiers have been demonstrated (Figure 27) from HF 74 to L band 75. Transmitters based upon the Kahn technique gen

48、erally have excel- lent linearity because linearity depends upon the modulator rather than RF power transistors. The two most important factors affecting the linearity are the envelope bandwidth and alignment of the envelope and phase modulations. As a rule of thumb, the envelope bandwidth must be a

49、t least twice the RF bandwidth and the misalignment must not exceed one tenth of the inverse of the RF bandwidth 76. In practice, the drive is not hard-limited as in the classical implementation.Drive power is conserved by allowing the drive to follow the envelope except at low levels. The use of a minimum drive level ensures proper operation of the RF PA at low signal levels where the gain is low 77. At higher microwave frequencies, the RF power devices exhibit softer saturation characteristics and larger amounts of amplitude-to-phase conversion, necessitating the use of predistort